1. Field of the Invention
The present invention relates to a switching power supply circuit.
2. Description of the Related Art
As so-called soft-switching power supply of a resonant type, a voltage resonant converter formed by a single-ended system with one switching element as in Japanese Patent Laid-Open No. 2000-134925, for example, is known.
FIG. 22 shows one example of configuration of a switching power supply circuit including a voltage resonant converter formed by such a single-ended system.
The switching power supply circuit shown in FIG. 22 rectifies and smoothes an alternating voltage VAC from a commercial alternating-current power supply AC by a rectifying and smoothing circuit composed of a bridge rectifier circuit Di and a smoothing capacitor Ci, and thereby generates a rectified and smoothed voltage Ei as voltage across the smoothing capacitor Ci. Incidentally, a noise filter, which is composed of one set of common mode choke coils CMC and two across capacitors CL, for eliminating common mode noise is provided in the line of the commercial alternating-current power supply AC.
This rectified and smoothed voltage Ei is input as direct-current input voltage to the voltage resonant converter. This voltage resonant converter has a configuration of a single-ended system having one switching element Q1. The voltage resonant converter in this case is externally excited, and thus a MOS-FET as the switching element Q1 is switching-driven by an oscillating and driving circuit 2.
A body diode DD of the MOS-FET is connected in parallel with the switching element Q1. In addition, a primary side parallel resonant capacitor Cr is connected in parallel with the drain and source of the switching element Q1.
A primary side parallel resonant circuit (voltage resonant circuit) whose resonance frequency is governed by the primary side parallel resonant capacitor Cr and a leakage inductance L1 produced by the primary winding N1 of a converter transformer PIT is formed. A voltage resonant operation as switching operation of the switching element Q1 is obtained by the primary side parallel resonant circuit. The resonance frequency is determined mainly by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side parallel resonant capacitor Cr.
The oscillating and driving circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1 to switching-drive the switching element Q1. Thereby the switching element Q1 performs switching operation at a switching frequency corresponding to the cycle of the drive signal.
The converter transformer PIT transmits the switching output of the switching element Q1 to a secondary side. The structure of the converter transformer PIT has for example an EE-shaped core formed by combining E-shaped cores of ferrite material with each other. A winding part is divided between the primary winding N1 and a secondary winding N2, and the primary winding N1 and the secondary winding N2 are wound on a bobbin around a central magnetic leg of the EE-shaped core. Also, a gap of 0.8 mm to 1 mm, for example, is formed in the central magnetic leg of the EE-shaped core of the converter transformer PIT. Thereby k=about 0.80 to 0.85 is obtained as the value of a coupling coefficient k between the primary side and the secondary side. The coupling coefficient k at this level represents a degree of coupling that may be considered to be loose coupling. Lowering the value of the coupling coefficient k makes the converter transformer saturated less easily. In addition, the leakage inductance L1 is produced in the primary winding N1 on a condition that the value of the coupling coefficient k is lower than one.
One end of the primary winding N1 of the converter transformer PIT is inserted between the switching device Q1 and the positive electrode terminal of the smoothing capacitor Ci. Thereby, the switching output of the switching device Q1 is transmitted to the primary winding N1. An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 of the converter transformer PIT.
In this case, a secondary side series resonant capacitor C5 is connected in series with one end of the secondary winding N2. Thereby a secondary side series resonant circuit (current resonant circuit) whose resonance frequency is governed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C5 is formed. The resonance frequency is determined mainly by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C5.
In addition, rectifier diodes Do1 and Do2 and a smoothing capacitor Co are connected to the secondary side series resonant circuit as shown in FIG. 22, whereby a voltage doubler half-wave rectifier circuit is formed. This voltage doubler half-wave rectifier circuit generates a secondary side direct-current output voltage Eo having a level corresponding to twice an alternating voltage V22 induced in the secondary winding N2 as voltage across the smoothing capacitor Co. The secondary side direct-current output voltage Eo is supplied to a load, and is input to a control circuit 1 as detection voltage for constant-voltage control.
The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side direct-current output voltage Eo input as detection voltage to the oscillating and driving circuit 2. The oscillating and driving circuit 2 outputs a drive signal varied in frequency and the like according to the level of the secondary side direct-current output voltage Eo which level is indicated by the detection output input to the oscillating and driving circuit 2. The oscillating and driving circuit 2 thereby controls the switching operation of the switching element Q1 so as to make the secondary side direct-current output voltage Eo constant at a predetermined level. Thereby control for stabilizing the secondary side direct-current output voltage Eo is performed.
FIGS. 23A, 23B, and 23C and FIG. 24 show results of experiments on the power supply circuit having the configuration shown in FIG. 22 described above. Incidentally, in the experiments that provided the results of FIGS. 23A, 23B, and 23C and FIG. 24, principal parts of the power supply circuit of FIG. 22 is set as follows.
For the converter transformer PIT, an EER-35 is selected as a core material, and the gap of the central magnetic leg is set to a gap length of 1 mm. As for the numbers T of turns of the primary winding N1 and the secondary winding N2, N1=39 T and N2=23 T, respectively. The level of a voltage induced per turn (T) of the secondary winding N2 is set to 3 V/T. The coupling coefficient k of the converter transformer PIT is set to k=0.81.
The capacitance of the primary side parallel resonant capacitor Cr is selected to be Cr=3900 pF, and the capacitance of the secondary side series resonant capacitor C5 is selected to be C5=0.1 μF. Accordingly, the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit is set at 230 kHz, and the secondary side series resonance frequency fo2s of the secondary side series resonant circuit is set to 82 kHz. In this case, relative relation between the primary side parallel resonance frequency fo1p and the secondary side series resonance frequency fo2s is expressed by fo1p≈2.6×fo2s.
The rated level of the secondary side direct-current output voltage Eo is 135 V. Load power is supplied in a range of maximum load power Pomax=200 W to minimum load power Pomin=0 W.
FIGS. 23A, 23B, and 23C are waveform charts showing operations of principal parts in the power supply circuit shown in FIG. 22 on the basis of switching cycles of the switching element Q1. FIG. 23A shows a voltage V1, a switching current IQ1, a primary winding current I21, a secondary winding current I22, and secondary side rectified currents ID1 and ID2 when the maximum load power Pomax=200 W. FIG. 23B shows the voltage V1, the switching current IQ1, the primary winding current I21, and the secondary winding current I22 when medium load power Po=120 W. FIG. 23C shows the voltage V1 and the switching current IQ1 when the minimum load power Pomin=0 W.
The voltage V1 is obtained across the switching device Q1. The voltage V1 is at a zero level in a period TON in which the switching device Q1 is on, and forms a sinusoidal resonant pulse in a period TOFF in which the switching device Q1 is off. The resonant pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is voltage resonant operation.
The switching current IQ1 flows through the switching device Q1 and the body diode DD. The switching current IQ1 flows with waveforms shown in the figures during the period TON, and is at a zero level during the period TOFF. The primary winding current I21 flowing through the primary winding N1 is obtained by combining a current component flowing as the switching current IQ1 during the period TON with a current flowing through the primary side parallel resonant capacitor Cr during the period TOFF.
The rectified currents ID1 and ID2 shown in FIG. 23A flowing through the rectifier diodes Do1 and Do2 forming a secondary side rectifier circuit, each flows sinusoidally, as shown in the figure. In this case, the resonant operation of the secondary side series resonant circuit appears in the waveform of the rectified current ID1 more dominantly than in the rectified current ID2.
The secondary winding current I22 flowing through the secondary winding N2 has a waveform obtained by combining the rectified currents ID1 and ID2 with each other.
FIG. 24 shows switching frequency fs, the on period TON and the off period TOFF of the switching element Q1, and AC-to-DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 22 with respect to load variation.
First, looking at the AC-to-DC power conversion efficiency (ηAC→DC), high efficiencies of 90% or more are obtained over a wide range of load power Po=50 W to 200 W. The inventor of the present application has previously confirmed by experiment that such a characteristic is obtained when a secondary side series resonant circuit is combined with a voltage resonant converter of a single-ended type.
The switching frequency fs, the on period TON, and the off period TOFF in FIG. 24 indicate the switching operation of the power supply circuit shown in FIG. 22 as a characteristic of constant-voltage control dealing with load variation. In this case, the switching frequency fs is substantially constant with respect to load variation. On the other hand, the on period TON and the off period TOFF linearly change in manners opposite to each other as shown in FIG. 24. This indicates that the switching operation is controlled by holding the switching frequency substantially constant while the secondary side direct-current output voltage Eo is varied, and changing a time ratio between the on period and the off period. Such control can be regarded as PWM (Pulse Width Modulation) control that changes the on period and the off period within one cycle. The power supply circuit shown in FIG. 22 stabilizes the secondary side direct-current output voltage Eo by this PWM control.
FIG. 25 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 22 by relation between the switching frequency fs (kHz) and the secondary side direct-current output voltage Eo. The power supply circuit shown in FIG. 22 has the primary side parallel resonant circuit and the secondary side series resonant circuit. Therefore the power supply circuit shown in FIG. 22 has, in a composite manner, two resonant impedance characteristics, that is, a resonant impedance characteristic corresponding to the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit and the secondary side series resonance frequency fo2s of the secondary side series resonant circuit. Since the power supply circuit shown in FIG. 22 has the relation fo1p≈2.8×fo2s, the secondary side series resonance frequency fo2s is lower than the primary side parallel resonance frequency fo1p, as shown in FIG. 25.
As for constant-voltage control characteristics with respect to the switching frequency fs under a condition of a constant alternating input voltage VAC, as shown in FIG. 25, constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit are represented by characteristic curves A and B, respectively, and constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the secondary side series resonance frequency fo2s of the secondary side series resonant circuit are represented by characteristic curves C and D, respectively. When constant-voltage control is to be performed at tg, which is a rated level of the secondary side direct-current output voltage Eo, under the characteristics shown in FIG. 25, the variable range (necessary control range) of the switching frequency fs necessary for the constant-voltage control can be represented as a section denoted by Δfs.
The variable range Δfs shown in FIG. 25 as a necessary frequency control range extends from the characteristic curve C, which corresponds to the secondary side series resonance frequency fo2s of the secondary side series resonant circuit at the maximum load power Pomax, to the characteristic curve B, which corresponds to the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit, at the minimum load power Pomin. Crossed between the characteristic curve C at the maximum load power Pomax and the characteristic curve B at the minimum load power Pomin are the characteristic curve D, which corresponds to the secondary side series resonance frequency fo2s of the secondary side series resonant circuit, at the minimum load power Pomin and the characteristic curve A, which corresponds to the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit, at the maximum load power Pomax. Thus, as the constant-voltage control operation of the power supply circuit shown in FIG. 22, switching-driving control is performed by PWM control that holds the switching frequency fs substantially fixed, and which changes the time ratio between the periods TON and TOFF in one switching period. Incidentally, this is indicated by the fact that the period length of one switching period (TOFF+TON) shown at the times of the maximum load power Pomax=200 W, load power=100 W, and the minimum load power Pomin=0 W in FIGS. 23A, 23B, and 23C is substantially constant, while the widths of the period TOFF and the period TON are changed.
Such an operation is obtained by making a transition between a state in which the resonant impedance (capacitive impedance) at the primary side parallel resonance frequency fo1p of the primary side parallel resonant circuit is dominant and a state in which the resonant impedance (inductive impedance) at the secondary side series resonance frequency fo2s of the secondary side series resonant circuit is dominant, as resonant impedance characteristics according to load variation in the power supply circuit, in the narrow variable range (Δfs) of the switching frequency.
The power supply circuit shown in FIG. 22 has the following problems. The switching current IQ1 at the maximum load power Pomax shown in FIG. 23A among the waveform charts of FIGS. 23A, 23B, and 23C described above is at a zero level until an end of the off period TOFF as turn-on timing. When the on period TON arrives, the switching current IQ1 operates such that the switching current IQ1 first flows as a current of negative polarity through the body diode DD1 and is then inverted to flow between the drain and the source of the switching element Q1. Such an operation indicates that ZVS (Zero Voltage Switching) is performed properly.
On the other hand, the switching current IQ1 shown in FIG. 23B when Po=120 W corresponding to a medium load operates so as to flow as noise before the end of the off period TOFF as turn-on timing. This operation is an abnormal operation in which ZVS is not properly performed. That is, it is known that a voltage resonant converter provided with a secondary side series resonant circuit as shown in FIG. 22 causes the abnormal operation in which ZVS is not properly performed at the time of the medium load. It is confirmed that an actual power supply circuit as shown in FIG. 22 causes such an abnormal operation in a load variation range as the section A shown in FIG. 24, for example.
As described earlier, a voltage resonant converter provided with a secondary side series resonant circuit inherently has, as a tendency thereof, a characteristic of being able to maintain high efficiency in an excellent manner as the load is varied. However, as shown by the switching current IQ1 in FIG. 23B, a considerable peak current flows at the time of turning on the switching element Q1, thereby inviting an increase in switching loss and constituting a factor in lowering power conversion efficiency. At any rate, the abnormal operation as described above causes a shift in phase-gain characteristics of a constant-voltage control circuit system, for example, thus resulting in a switching operation in a state of abnormal oscillation. Thus, in the present situation, there is a strong recognition that it is difficult to put the power supply circuit of FIG. 22 to practical use in actuality.
In order to remedy this, a so-called class E switching converter shown in FIG. 26 has been put to practical use as a converter combining a voltage resonant converter and a current resonant converter. The class E switching converter shown in FIG. 26 has a switching element Q1. The switching element Q1 in this case is a MOS-FET. A body diode DD is formed so as to be connected in parallel with the drain and source of the switching element Q1 as MOS-FET. A forward direction of the body diode DD in this case is from the source to the drain.
A primary side parallel resonant capacitor Cr is connected in parallel with the drain and source of the same switching element Q1. The drain of the switching element Q1 is connected to the positive electrode terminal of a direct-current input voltage Ein via a series connection of a choke coil L10. The source of the switching element Q1 is connected to the negative electrode terminal of the direct-current input voltage Ein.
One terminal of a choke coil L11 is connected to the drain of the switching element Q1, and a series resonant capacitor C11 is connected in series with another terminal of the choke coil L11. An impedance Z as a load is inserted between the series resonant capacitor C11 and the negative electrode terminal of the direct-current input voltage Ein. Concrete examples of the impedance Z include a piezoelectric transformer, a fluorescent lamp ready for high frequencies, and the like.
Incidentally, since the inductance of the choke coil L10 is set considerably higher than the inductance of the choke coil L11, the class E switching converter of such a configuration can be considered one form of a complex resonant converter including a parallel resonant circuit formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonant capacitor Cr and a series resonant circuit formed by the inductance of the choke coil L11 and the capacitance of the series resonant capacitor C11. In addition, the class E switching converter can be said to be the same as a voltage resonant converter of a single-ended type in that the class E switching converter is formed with only one switching element.
FIG. 27 shows operations of principal parts of the class E switching converter having the configuration shown in FIG. 26 described above. A switching voltage V1 is obtained across the switching element Q1. The switching voltage V1 is at a zero level during a period TON during which the switching element Q1 is on, and has a sinusoidal pulse waveform during a period TOFF during which the switching element Q1 is off. This switching pulse waveform is obtained as a result of the resonant operation (voltage resonant operation) of the above-described parallel resonant circuit.
A switching current IQ1 flows through the switching element Q1 and the body diode DD. The switching current IQ1 is at a zero level during the period TOFF. In the period TON, the switching current IQ1 first flows through the body diode DD and is thus of negative polarity for a certain period from a starting time point of the period TON, and is thereafter inverted to be of positive polarity and flows from the drain to the source of the switching element Q1.
A current I2 flowing through a transformer of the class E switching converter is obtained by combining the switching current IQ1 flowing through the switching element Q1 (and the body diode DD) with a current flowing through the primary side parallel resonant capacitor Cr. The current I2 has a waveform including a sinusoidal wave component.
Relation between the switching current IQ1 and the switching voltage V1 described above indicates that the operation of a ZVS characteristic is obtained in turn-off timing of the switching element Q1 and that the operation of the ZVS characteristic and a ZCS (Zero Current Switching) characteristic is obtained in turn-on timing.
A current I1 flowing from the positive electrode terminal of the direct-current input voltage Ei through the choke coil L10 into the class E switching converter forms a pulsating current waveform having a predetermined average level as shown in FIG. 27 with a relation of L10>L11 set for the inductances of the choke coils L10 and L11. Such a pulsating current waveform can be considered approximate to a direct current.